Steerable antenna array



March l, 1966 Q F, vQG-r STEERABLE ANTENNA ARRAY 6 Sheets-Sheet 1 Filed NOV. 28, 1962 c. ...Zus-mju 20mm ATTORNEK March 1, 1966 G. F. voGT STEERABLE ANTENNA ARRAY 6 Sheets-Sheet 2 Filed Nov. 28, 1962 ATTORNEY.

March 1, 1966 Filed Nov. 28, 1962 FIG. 8

6 Sheets-Sheet 5 ffn coMPARlsoN-l 2m vez n RU|T II L CQNTRO'- m L o6 ml. IcoNTRoL ELEMENT /voLTAGE I PULSE FuNc-nou lO n. \F il SHAPE@ GENERAToR m y To com'. Ro'cl SAWTOOTH .tf-|04 SmJF/AFSEER 402 GENERAroR I CCOO `LI oz FREouENcvf' GENERATOR F G. 3 Trn BA 1x B Vc t, fn 12 t3 7 V l VC/ s 1 1 AUXLARY i E l l FREQUENCY l l l A D D ff GATING PULSE rCP r a fn` 1 F lf2 RoTATloN an I K E u E n o w in@ CONTROL FUNCTION ik F INVENTOR. a GOTTFR/ED F. VOGT G tvb" Bl/w? M TC ATTORNEY.

G. F. VOGT STEERABLE ANTENNA ARRAY INVENTOR, GOTTFR/ED F.' VOGT.

(man m mm i March 1, 1966 l Filed Nov. 28, 1962 ...ani/ E 0 m .mi

ATTORNEY March l, 1966 G. F. voGT STEERABLE ANTENNA ARRAY 6 Sheets-Sheet 5 Filed Nov. 28, 1962 vom @02x wom\ NOM QOS- ATTORNEY.

March l, 1966 Filed Nov. 28, 1962 FIG. Il

G. F. voGT 3,238,527

STEERABLE ANTENNA ARRAY 6 Sheets-Sheet 6 INVENTOR, GoTTFmEo F. vom

ATTOR NEYJS` United States Patent O 3,238,527 STEERABLE ANTENNA ARRAY Gottfried F. Vogt, Lincroft, NJ., assigner to the United Strates of America as represented by the Secretary of the Filed Nov. 28, 1962, Ser. No. 240,762 l5 Claims. (Ci. 343-100) (Granted under Title 35, US. Code (1952), sec. 266) The invention described herein may be manufactured and used by or for the Government for governmental purposes Without the payment of any royalty thereon.

This invention relates to Steerable antenna arrays and more particularly to a Steerable antenna array adapted to be steered by electronic means.

Over the past three decades, antenna arrays with a Steerable antenna pattern have been designed and operated. The main purpose of these arrays has been to combat multipath effects and undesired signals in the high frequency (HF) range. One of the best known examples of linear array type Steerable antennas is employed by the MUSA (Multiple Unit Steerable Antenna) system. However, such systems Were not employed more often because of their enormous size which made the cost prohibitive.

Since narrow beam Widths of only a few degrees combined with a relatively low frequency range require an aperture in the order of several miles, it is evident that the beam of such antennas cannot be controlled by mechanical motion of the entire array as is done With radar antennas at high frequencies. The steering of the vertical angle of incidence is generally accomplished by individual phasing of the antenna element channels with successive combining. In conventional steering arrays now utilized, the phasing and receiving equipment are located in a control center and the individual elements are connected by cables to the -control center. These systems, which feature a common phasing center for all elements of the array, require expensive cabling from the individual antenna element to the phasing center. Moreover, the phase Shifters in such systems are usually of the electromechanical type and perform a continuous phase shift of n times 360 by employing the capacitive or the inductive principle and, accordingly, the magnitude of phase shift necessary to form and control the beam is performed by mechanical gear drives.

Other principles of phasing may employ mechanically switched delay lines. An example of this application is the Wullenweber antenna, which allows control of the pattern in the azimuthal plane and the ISCAN (lnertialess Steerable Communications Antenna) system. While the broadband frequency characteristic attained with such systems is quite desirable, it has been found that, with respect to cable costs and to some extent the complexity of the switching circuitry, the delay line principle seriously limits any further developments having tighter pattern requirements. Furthermore, such systems require a central steering matrix and can not be remotely controlled.

It is an object of the present invention to provide an improved Steerable antenna array wherein the above limitations are overcome.

It is another object of the present invention to provide an improved Steerable antenna array wherein the system is completely electronically controlled thus eliminating the need of mechanically rotating parts.

It is another object of the present invention to provide an improved Steerable antenna array wherein the number of cables and concomitant cabling costs are drastically reduced.

In accordance With one embodiment of the present in- Vention, there is provided a Steerable antenna system which includes a plurality of antenna elements spaced with respect to a reference antenna element such that a wave front of hgh frequency radio energy at an antenna element is at a prescribed time phase with respect to the wave front of said high frequency radio energy at the reference antenna. Included further are means for generating a signal voltage having a magnitude proportional to the angle of incidence of the wave front of the high frequency radio energy and discrete phase Shifters in circuit with respective antenna elements except the reference antenna element, the phase shifters being responsive to the high frequency radio energy. Also included are discrete means respectively in circuit with the phase Shifters and responsive to said signal voltage whereby the high frequency radio energy at respective phase Shifters is shifted in phase by an amount corresponding to the time phase of the high frequency radio energy at the associated antenna element. Also included are respective adder circuits responsive to the radio frequency energy outputs of respective phase shifters and to the radio frequency energy output of the reference antenna element. The adder circuits are serially connected such that the output of any one phase shifter is added to the outputs of preceding phase Shifters and the combined added phase shifter outputs are added to the output of the reference antenna element.

In accordance with another embodiment of the invention, there is provided a system for electronically scanning and orienting the angle of linear polarization of electromagnetic incident Waves. It includes a pair of quadrature positioned crossed dipole antennas, one of the antennas being positioned along a prescribed horizontal axis such that the vector representing the incident Wave signal forms a prescribed angle of polarization with respect to the horizontal axis. This provides hor-izontal and vertical volta-ge signal components of the incident signal vector which are cosine and sine functions, respectively, of the angle of polarization, and which are developed across the horizontal and vertical dipoles, respect-ively. Included further is a lirst and second capacitive bridge type modulator which includes means responsive, respectively to said horizontal and vertical voltage signal components and means for additively combining the outputs of t-he modulators. Also included are means for generating a signal voltage having a magnitude proportional to the angle of polarization of the incident wave signal, and means for sweeping the signal voltage in accordance with a scanning signal. Further included are means in circuit with the capacitive bridge modulators and responsive to the swept signal voltage for simultaneously generating a pair of signal voltages which are sine and cosine functions, respectively, of the polarization angle. The signal voltage sine function and the horizontal voltage signal component are multiplied in one of the modulators, and the signal voltage cosine function and the vertical signal component are multiplied in the second modulator. The combined output of the modulators is the sum of the multiplied signals. Further ineluded is a receiver responsive to the sum signal from the modulators, the receiver being adapted to produce IFv signals. Also included is an oscilloscope having vertical and horizontal deflection plates with the vertical deflection plates being responsive to the IF receiver signal and the horizontal deflection plates being responsive to the scanning signals `such that the pattern formed on the face of the oscilloscope is an indication of the polarization angle.

For a better understanding of the invention, together with other and further objects thereof, reference is had to the following description taken in connection with the accompanying drawings in which:

FIG. 1 is a schematic diagram of a Steerable antenna system in accordance with the invent-ion;

FIG. 2 is a block schematic diagram of an electronical- Patented Mar. 1, 1966l 3 ly controlled phase shifter associated with antenna elements of FIG. 1;

FIGS. 3, 4, 9 and 10A are explanatory curves;

FIG. 5 is a schematic representation of a non-uniformly distributed antenna array;

FIG. 6 is a detailed schematic diagram of the phase shifter circuit;

FIG. 7 is a detailed circuit diagram of the element function generator;

FIG. 8 is a schematic diagram illustrating the various modes of operation of the steerable antenna system of FIG. 1;

FIG. 10 is a schematic diagram illustrating a second embodiment of the invention wherein the angle of linear polarization may be electronically scanned and/or orientated; and

FIG. ll is a schematic block diagram illustrating the means for generating a control voltage signal in the system of FIG. 1.

Referring now to FIG. l of the drawing, there is shown at an endfire antenna array including ru elements, which may be either uniformly or non-uniformly dis-- tributed, numbered consecutively 0, l, 2, 3, n. The respective outputs of each of the antenna elements following the 0 element are channeled to respective adder circuits 2 through respectively amplifiers A and respective continuous electronically controlled phase shifters PH, which, in turn, are controlled by respective element function generators EFG. These elements are shown as blocks with corresponding subscripts to indicate the associated antenna element. A complete description of these blocks will be presented below. The output of the 0-element antenna is channeled directly to its respective adder E0 through amplifier A0. As shown, each of the element function generators EFG is controlled by a common signal derived from a control signal converter CSC, which in turn is controlled by a control signal CS, which, as hereinafter described, provides a signal which is a function of the magnitude of the angle of arrival gb of a wavefront. The respective outputs of the phase shifter elements PH, and antenna element 0, are applied to the respective associated adder circuits 2 which are, in turn, interconnected by a common cable CA such that the output of any one phase shifter element PH is added to the outputs of preceding phase Shifters and the combined added phase shifter outputs are added to the output of antenna element 0. The adder E0 corresponding to antenna element 0, hereinafter referred to as the reference element, delivers the total HF energy to a receiver RX. This energy is the sum of all vectors generated by the n+1 antenna elements. The time phase of a signal received at an antenna element with respect to the reference antenna element 0 may be expressed by the well known equation mil e l Vk p cos Xm=distance between antenna reference element 0 and antenna element n fH=vfrequency of operation Vk=velocity of propagation in space where V0=velocity of propagation in cable CA p=angle of vertical incidence of the signal wavefront.

in such communication systems, it may be required to provide continuous beam rotation from zero to 360. These three modes may be respectively defined as (l) the scanning mode, (2) the orientation mode, and (3) the continuous rotation mode. As hereinafter explained, these three modes may be readily achieved electronically by the antenna system illustrated in FIG. 1 where respective phase Shifters are controlled by associated function generators, which in turn are controlled from a common input information. These elements will now be described in detail.

FIG. 2 illustrates in block diagram form the functional diagram of the control center CC and an electronically controlled phase shifter -for one antenna element in accordance with the invention. It is to be understood that the desired output from a respective phase shifter PHn is a phase angle n such that at each antenna element n,

In order to facilitate the explanation of the circuit shown in FIG. 2, it will be assumed that the orientation mode of operation is required, i.e., the antenna beam is to be positioned at a prescribed vertical incidence angle and maintained in this position for an indefinite time.

CONTROL CENTER CC (100) Referring now to FIGS. 2 and 3, the control center CC includes a master or clock frequency generator 1.02 for generating a reference sine-wave signal at frequency fm, FIG. 3A, from which are derived a train of reference pulses having a repetition rate fm. The time between successive reference pulses is therefore as shown in FIG. 3B. The reference pulses at repetition rate fm are applied to a sawtooth generator circuit 104 from which there is derived a periodic sawtooth voltage signal at a repetition rate of fm and having a peak voltage VS as shown in FIG. 3C. The sawtooth voltage is applied as one input to a comparison circuit 106, the other input to comparison circuit 106 being a D.C. control voltage signal V,3 derived from source 108 whose magnitude is a function of the angle of vertical incidence qb and hence is proportional to the input control information. The output of comparison circuit 106 provides a control pulse CP when the amplitude of the sawtooth voltage and Vc are equal. With such an arrangement the control signal is converted to a control pulse CP generated at the repetition rate fm, the pulse CP occurring with a time delay Tc after the starting reference pulse so that, in effect, Tc contains the control information in the form of a pulse position modulation. The comparison circuit input is shown in FIG. 3C and the control pulse CP in relationship to the reference pulses is shown in FIG. 3D. The control pulses CP and the clock generator frequency fm are utilized as contr-ol signals for each of the element function generators which in turn control the output of the respective phase Shifters PH as explained below.

ELEMENT FUNCTION GENERATOR EFG (200) The function generator includes two auxiliary frequency generators 206 and 208 which are adapted to provide the same frequency oscillation outputs fn, but differing in phase by so that one frequency generator, 206 for example, provides a sine-wave function fn 0 and the other frequency generator, 208, provides a cosinewave function fn 90. This relationship is shown in FIG. 3E. As hereinbelow explained, the particular frcquency fn for a prescribed element function generator is a function of (l) clock frequency fm; (2) the distance Xn between the associated particular antenna element and the reference antenna element, which for this application is assumed to be at 0, and (3) the operating frequency fH. Auxiliary frequency fn may be harmonically related to the clock frequency fm for an endre array having uniform element distribution, but is not so related for an endre array having a non-uniform element distribution. However, in both cases, it is required that the start of auxiliary sine wave oscillation fn be conicident with the start of each cycle of clock frequency fm, i.e., at the zero crossing of fm at amplitude zero. This can best be seen in FIG. 4 where t represents the start of a cycle of clock frequency fm and t represents the end of the cycle. For a uniformly distributed endiire array, the auxiliary frequency is shown as f1 which is harmonically related to the clock frequency fm. For a non-uniformly distributed endiire array as shown in FIG. 5, -for example, the auxiliary frequency is shown at f2. In order for f2 to start at the zero crossing of clock frequency fm with zero amplitude, the wave f2 must be damped at t" as shown. It can be seen that for the harmonically related oscillation fn, the zero crossings of fm and fn coincide at t and t and no damping is required.

Each of the respective oscillation signals derived from generators 206 and 208 are fed torespective boxcar generator circuits 210 and 212 through respective gating circuits 214 and 216. The gating circuits 214 and 216 are simultaneously activated only at time Tc by the control pulse CP (FIG. 3F) which may be narrowed by passing it through a control pulse shaper 218. Gating circuits 214 and 216 are preferably of the fast-acting, low impedance type well known in the art. The respective outputs of gates 214 and 216 are filtered by the respective boxcar circuits 210 and 212 to provide D.C. voltages whose respective magnitudes are proportional to the instantaneous amplitude of the respective signals fn 0 and f5, 90 at time Tc. As is well known in the art, these D.C. voltages will be held by boxcar circuits 210 and 212 until other amplitudes of signals fn 0 and fn 90 are passed through respective gates 214 and 216. Designating the voltage amplitude of the gated sine wave signal fn 0 as an and a voltage amplitude of the wave signal f5, 90 as bn, then it can be seen that the magnitude of an and b5, in the rst order is a D.C. voltage. Thus the two control voltages derived from the boxcar circuits 210 and 212 may be expressed as bn=K cos ZWTCfn (4) where T5=time delay of control pulse CP These control voltages are shown in FIG. 3G. The relationship between Tc and Tm may be expressed as and since, from (4), the phase angle of the control information is n=21r Tc fn, it follows from Equation 7 that 6 Substituting Equation 8 in Equations 3 and 4 we have KSL-) f ctn-K sin (2n-Vs fm Ir sul n (9) and .1 bn-K cos (21rVs fm K eos ,Bn (10) The control signals of Equations 3 and 4 are generated or produced by the element function generator EFG and are supplied as respective inputs to the phase shifter as hereinafter explained.

The detailed schematic diagram of an element function generator is illustrated in FIG. 7. For ease of explanation, the auxiliary frequency fn is assumed to be the second harmonic of the clock frequency fm, i.e., fnz2fm. Referring now to FIG. 7, the clock frequency fm is shown applied to a conventional frequency doubler circuit 221, the output of which is simultaneously coupled to two tuned circuits 223 and 225 through isolation stage 227 and respective coupling capacitors 229 and 231. The coupling capacitors 229 4and 231 have relatively small capacitance value so that the tuned circuits 223 and 225 are isolated from each other at the auxiliary frequency fn. The parameters of the tuned circuits 223 and 225 are chosen such that one tuned circuit lags by 45 and the other leads by 45, in phase to provide a total phase shift of The sine-wave output of each tuned circuit 223 and 225 is fed to respective sampling gates 233 and 235. Each sampling lgate -consists of four fast switch diodes D5 which are connected to form a bridge network. As shown diodes D51 `and D52, and diodes D53 and D54, of sampling gate 233 are serially connected in the forward direction to provide two bridge arms. The cathode electrodes of D51 and D53 are interconnected 4by a balancing potentiometer 237 and the anode electrodes of D52 and D54 are interconnected by a balancing potentiometer 239. Similarly, diodes D55 and D56, and diodes D57 and D58, of sampling gate 235 are serially connected in the forward direction to provide two bridge arms. The cathode electrodes of D55 and D5, are interconnected by a balancing potentiometer 241 and, similarly, the anode electrodes of D56 and D58 are interconnected by a balancing potentiometer 243. The respective center arms of :balancing potentiometers 237 and 239 are interconnected by a circuit which includes one secondary winding 245 of a pulse transformer 247 in series connection with a capacitor 249. A Zener diode 251 is connected across capacitor 249 and a damping diode 253 is connected across winding 245, the diodes being poled in opposite direction and the junction of the anode electrodes thereof being connected to the junction of capacitor 249 and winding 245. Similarly, the respective center arms of balancing potentiometers 241 and 243 are interconnected by a circuit which includes another secondary winding 255 of pulse transformer 247 in series connection with a capacitor 257. A Zener diode 259 is connected across capacitor 257 and a damping diode 261 is connected across winding 255. By such an arrangement the windings 245 and 255 may be considered to be two floating secondary windings. The control pulse CP is applied to the primary winding 263 of transformer 247 through an Avalanche Breakdown Pulse Shaper 265 -well known in the art. The pulse transformer 247 thus feeds the two sampling gates 233 and 235 simultaneously yby means of the two iioating secondary windings hereinabove described. The remaining two bridge points of sampling gate 233 at the respective junctions of switch diodes D51 and D52, and D53 and D54 serve, respectively, to accept the continuous frequency fn-45" from tuned circuit 223 and to deliver the depicted instantaneous `amplitude across a capacitor 267 connected between the junction of D53 and D54, and ground. Similarly, the remaining two bridge points of sampling gate 235 at the respective junctions `of switch diodes D55 and D56, and D57 and D58 serve, respectively, to accept the continuous frequency n-{45 from tuned circuit 225 and to deliver the depicted instantaneous amplitude across a capacitor 269 connected between the junction of D57 and D53 and ground. The capacitors 267 and 269 operate as the boxcar filters. In order to operate as desired, the capacitors of tuned circuits 223 and 225 must be large compared to the boxcar capacitors 267 and 269. Since all the switching diodes D51-D58 are in the forward direction at the same time, the pulse occurs through transformer 247 and is :balanced by means of the balancing potentiometers 237-243. The control pulse, CP, itself will not generate any -output Voltage across capacitors 267 and 269. The voltage `across capacitors 249 and 257 is generated by the pulses derived from pulse transformer 247. Effectively, both capacitors 249 and 257 are substantially charged to the peak value of the control pulse CP. The Zener diodes 251 and 261 operate to stablize the voltage across respective capacitors 249 and 257 to a predetermined value, preferably 3 volts. Between control pulses, the diodes D51-D54 of gate 233, and diodes D55-D58 of gate 235, are reversed Ibiased by the voltages `on respective capacitors 249 and 257 so that between control pulses, CP, no instantaneous voltage is applied from the tuned circuits 223 and 225 to affect the outputs of boxcar capacitors 267 and 269. When the diodes D5 are rendered conductive, a low impedance path is presented from the tuned circuits 223 and 225 to the boxcar capacitors 267 and 269 for the narrow duration of the control pulses CP. The output voltage across capacitors 267 and 269 will thus be able to swing to the positive `or negative side representing the true magnitude of the instantaneously depicted sine or cosine wave shapes of auxiliary frequency fn. The Zener diodes 251 and 259 enforce a stable bias voltage for the off position which is generated across the capacitors 249 and 257 by the control pulse CP during the conducting period of the switching diodes D51-D5s- The amplitude of the output voltage from the EFG is such that the modulators in the phase shifter PH operate in a linear fashion and provide sufficient gain.

PHASE SHIFTER PH (300) The fundamental principle of the phase shifter PH follows the simple trigonometric equation where HIHF input frequency at amplitude H Iphase angle information of amplitude K It can be seen from Equation l1 that the phase shifter output is obtained by the superposition, or adding, of two branches each of which respectively represents the product of the input signal multiplied with the control information derived from the element function generator EFG. As shown in FIG. 2, the phase shifter PH includes a modulator 302 to which is applied the control voltage a=K sin ,Bn and the high frequency input signal H sin ZWHI, and a modulator 304 to which is applied the control voltage b21( cos n and the high frequency input signal H cos 2 vrfHt. The mutiplied signal output from modulator 302 is accordingly (H sin 2f fHf) (K sin an) 12) and the multiplied output from modulator 304 is accordingly (H cos 21r fHt) (K cos n) (13) The signals of Equations 12 and 13 are combined in an adder 306 to provide the signal HK cos (21r fHt-n) (14) which represents the phase shifted high frequency HF output. Substituting the value of n in Equation 8 for [in in Equation 14, we have Equation 15 shows that the entire system functions as a phase shifter. From the above equations, it can be seen that the control information an=K sin n and bn=K cos n may be generated for a rate of angular change ,Bn in the audio frequency range down to D.-C., and the modulators operate as pure multipliers with modulation frequencies reaching from the audio range down to D.C. The A.C. requirement is a function of the scanning requirement of the steerable antenna beam, and the D.-C. requirement is fundamental for positioning the beam for an indefinite period, i.e., orientation mode operation.

The detailed schematic drawing of the phase shift element is shown in FIG. 6. Referring now to FIG. 6, the phase shifter includes two identically constructed modulators 302 and 304, hereinafter referred to as Modulator A and Modulator B, respectively, connected between the input and output of the phase shifter. The identical components of Modulators A and B are identified by the same reference numeral but are distinguished by the letter A or B following the reference numeral. Both modulators are of the carrier suppressed type. As shown, each modulator comprises a capacitive bridge network 311 connected across the input lines 313 and 315. Bridge networks 311 comprises two equal-valued fixed capacitors 317 and 319 and two voltage-variable capacitance diodes 321 and 323, i.e., voltage controlled varactors, connected across the lines 313 and 315, with the junction of the capacitors 317 and 319 being grounded as shown. The HF input signal 1HO is applied in push-pull to the input line 313A and 315A, and fH90 is applied in push-pull to input lines 313B and 315B by means of a conventional RC phase shifting network 325. By such an arrangement the input signal is fed in pushpull to the diagonal of bridge 311A and 311B. The voltage variable capacitors 321 and 323 are prebiased in the reverse direction by the D.C. voltage sources 327 and 329 through respective resistors 331 and 333. As shown, the biases are oppositely poled such that voltage variable capacitor 321 is negatively biased and capacitor 323 is positively biased with respect to the junction of the voltage sources 327 and 329 which is grounded as shown.

The control information L11-K sin is inserted singleended to modulator A through resistor 335A and the junction of the two voltage variable capacitors 321A and 323A. Similarly, control information b=K cos ,B is inserted single-ended to modulator B through resistor 335B and the junction of the two voltage variable capacitors 321B and 323B. The respective junctions of the Voltage variable capacitors provides the modulation product of the two modulator input signals and each modulation product is respectively coupled to a common output connection as at 337. The common output connection performs the function of the adder 306 shown in block form in FIG. 2. The gain of the modulators A and B are made equal by choosing the proper reverse biases for the voltage variable capacitance diodes 321, and they must be balanced to attain complete carrier suppression. With only fH applied to the modulator, the output therefrom at the junction of the voltage variable capacitors 321 and 323 is zero because the bridge is balanced. When control information is applied, the modulator becomes unbalanced since the voltage across one voltage variable capacitance, 321, increases the same amount that the other variable capacitance, 323, decreases. The modulator gain is therefore linear, and as a result pure multiplication is achieved. Since the higher side bands cancel out, only the lower side bands are added at the output 337 of the phase shifter.

It now remains to be shown how the values of the auxiliary frequencies fnl, fnz, etc., are derived and also the significance of the change in the pulse position TcfwVc.

Now, comparing Equation l where an=f(,Xn) (16) with Equation 8 where T D ..=f(T-m, (17) We can satisfy Equation 2 i.e. n=am by the following Hence, according to Equations 19 and 20 the following is the fundamental procedure for steering the beam of an endfire array.

(a) The angle of incidence is to be adjusted by a change of the pulse position T ,fel/C. This control information is the same for all individual phase Shifters` (b) The antenna element distance given in wavelengths determines the frequency ratio of the auxiliary frequency from generators 206 and 208 to the clock frequency derived from 102.

If, as in the conventional array the antenna element distance X1, X2 Xn followed a linear progression, the auxiliary frequencies f1, f2 fn are generated as harmonics of clock frequency fm as indicated in FIG. 4 where f1 is shown as the second harmonic of fm. However, if a non-uniform distribution tof the antenna elements is desired, the auxiliary frequencies need not be harmonically related. As hereinabove described in connection with FIG. 4, such a wave shape is given as f2 in FIG. 4. It is characteristic that this sine wave starts at the zero crossing of fm at the time t and must be damped at t in order to start again. It is evident that the controlfunction voltage bn (not shown in FIG. 4) starts with amplitude one at t'.

While the steerable antenna system described above has been shown as operating in the orientation mode inasmuch as Vc, the control voltage, is a D.C. voltage which is a function of the angle of incidence 0, this antenna system may be easily modified to operate in the scanning mode, and the continuous rotation mode. In the scanning mode, it will be necessary to scan the beam of the steeralble antenna back and forth in a limited range in order to distinguish the angle of incidence. This information will then be used to steer a second beam to this optimum position and remain there for a period of time, determined by the turbulence of the atmosphere. The modification for such a scanning mode merely comprises the utilization of a low frequency A.C. scanning Voltage Vc superimposed on Vc. The amplitude of Vc, of course, determines the scanning center and that of Vc the range of the scan. It is this superimposed voltage which is fed to the comparison circuit 106 of FIG. 2 so that the control pulse, CP, position is varied in accordance with the superimposed Vac. The control voltage Vc may be obtained by means of the system shownin FIG. 1l which is assumed to be operating in the scanning mode so that, in effect, there are two substantially identical systems using the same antenna structure. One system is to be operated as the orientation mode after the value of Vc has been determined by means of the scanning mode system. Referring now to FIG. 11, the HF beam is scanned by the output of Vac source 400 at, for example, 27 c.p.s. The `beam pattern from adder 306 of phase shifter 300, which represents the HF phase shifted signal, is applied to the vertical plates of oscilloscope 600. As s'hown, t-he -output of Vc 400 is applied as a sweep to the horizontal plates of oscilloscope 600. The maximum amplitude pattern of the incoming incident wave selected will appear to be off-center on the oscilloscope indicator as indicated by the dashed-line pattern. Since the initial setting -of Vc is such that the pattern of the selected incident wave is to be centrally positioned on the oscilloscope indicator, Vc is now varied until the maximum amplitude pattern (solid line) again appears at the center of oscilloscope 600. The magnitude of the Vc voltage required to perform this yoperation is thus proportional to the steering angle ,8. It is this VC value which may be manually or electronically fed to the comparison circuit 106 of the orientation mode operating system so that the magnitude of the Vc control voltage is now proportional to the angle of incidence of the wave front of the HF energy selected for communication.

The continuous rotation mode is utilized when the requirements demand a continuous beam rotation from zero to 360. To provide such a system, an additional oscillator is provided having the frequency rZJcmH-"Afs where fs is the scanning voltage Vc frequency and where determine the sense (clockwise or counterclockwise) and frequency of the rotation. In the continuous rotation mode, the auxiliary frequencies fn are generated as hereinabove described by the output of fm, but the control pulse CP will be generated in accordance With fr. That is, f, will pass through the pulse Shaper 218 of FIG. 2 and this Asignal at lfrequency fr will beat or move slowly against the auxiliary frequency fn thus depicting at every interaction a slightly different amplitude which generates the control function voltages. FIG. 8 illustrates schematically how these modes of operation may Ibe achieved by means of a ganged switch arrangement. The Vc source in FIG. 8 is shown at 400 and the additional oscillator for producing fr is shown at 402. The scanningj orientation and continuous rotation modes control the output of the element function generators in accordance with positions I, 1I and III, respectively, of ganged switch 404. The wave shape diagram of scanning mode switch position I, is shown in FIG. 9A, and the wave shape diagram of orientation mode switch position II is shown in FIG. 9B. FIGS. 9C and 9D illustrate the operation of the interaction of the .added oscillator frequency fr and the auxiliary frequency fn. FIG. 9E illustrates, on a compressed time scale, the different amplitude control function voltages derived from the interaction of fr, after being shaped by pulse Shaper 218, and the auxiliary frequency fn- F'IG. 10 illustrates a circuit wherein the control center 100, the element function generator 200, and the phase shifter 300 are applied to electronically rotate and orientate the linear polarization of an antenna and/ or to measure the polarization angle. Referring now to FIG. 10 where like elements are referred to by like reference numerals, the control center 100, the control pulse shaper 218, the element function generator 200, and the modulators 302 and 304 of phase shifter 300 are operationally and structurally identical to the like components hereinabove described. Also, the VC scanning source 400, the control Voltage source 108 and additionally fr frequency oscillator 402 are operationally and structurally identical to the like components hereinabove described, and the respective outputs of these elements are connected to the control shaper 218 and to comparison circuit 106 through ganged switch 404 as described inconnection with FIG. 8. In the circuit of FIG. 10, however, the output of control voltage source 108 is a D.C. control volt-age signal whose magnitude is a function of the angle of polarization, designated a, and hence is proportional to the input control information. der of the circuit of FIG. 10 will be described in connection with receiver operation for determining the polarization angle `of a received signal. As shown in FIG. 10, there is provided ,a pair of quadrature positioned coplanar crossed dipoles 504 and 506, one of which, 504 for example, may be considered to tbe horizontally polarized as designated by the X-axis. The other dipole 506 may therefore be considered to be vertically polarized as designated by the Y-axis. The dipoles employed may be any suit-able type well known in the art, such as. for instance, magnetic type loopstick antennas. It is to be noted that et is a spacial angle and that the time phase angle between t-he respective antenna terminals is for linear polarization. The respective output voltages Ex and Ey from antennas 504 and 506 represent two pushpull signals 90 out of phase. With the quadrature arranged antennas 5.04 and 506, Ex and Ey, the respective signal components of the antenna elements, are combined in such a way that the resultant plane of the electrical vector H of an incident wave forms the angle a with respect to the X-axis as shown in FIG. A. -By such an arrangement, X modulator 304 determines the amplitude and phase (0 or 180) of 'Ex, and the Y modulator 302 the amplitude and phase of Ey. The Ey signal is applied across capacitive bridge modulator 302 while the Ex signal is applied across capacitive bridge modulator 304.

The modulators 302 and 304 are controlled iby the output of element function generator 200 which, as previously described provides two sine wave signals 90 out of phase with respect to each other. In this case, however, the sine and cosine signals derived from element function generator 200 are functions of the polarizationangle a. Thus, the two wave signal outputs from element function generator 200 are designated A sin a and A cos a. These signals are generated by means of lche identical circuitry of control center 100 and element function generator 200 described in FIG. 2 and FIG. 7, taken in connection with FIG. 3. As described previously, the A sin a and A cos a control function signals are respectively applied to the junction of the variable capacitors in respective modulators 302 and 304. The multiplied outputs of modulators 302 and 304 are added as hereinabove described to produce a high frequency radio signal output equivalent to the function sin2 ot-I-cos2 a and constant vector amplitude may be maintained if the modulation process is made to satisfy the equation sin2 et-l-cos2 =l. Since the control function volt-age applied to the respective modulators 302 and 304 are 90 out of phase, lche polarization vector rotates through all four quadrants, i.e. from 0 to 360. Analogus to the phase shifter control circuit shown in FIG. 2, it can be shown that 2 T c.f n El. Tmfm- V. f...

where the functions on the rig-ht hand side are dened as indicated in Equations 5 through 8. To complete the circuit, the combined output from modulators 302 and 304 is fed to a receiver 506, whose IF output is applied to the vertical plates of an oscilloscope 508. The Vac output of generator 400 is applied to the horizontal plates of oscilloscope 508 as shown. Although the operation of the circuit shown in FIG. l0 is identical to that illustrated in FIG. 2, it would be advisable to illustrate the scanning mode of operation for position I of switch 404. In order to measure the angle a of the incident wave, the polarization of the antenna is swept by means of the scanning voltage Vc from source 400. The polarization angle in this particular example is swept back and forth starting from a -center position of 180J:l80. Thus, in order to measure the angle a of the incident wave, the polarization of the antenna is swept Iby means of the scanning generator 400. The rotation of the polarization characteristic Although not limited thereto, the remain-- iii will then produce a double sideband suppressed-carrier modulation of the incident carrier frequency, with the scanning frequency appearing as the modulation frequency. Using the IF output of the receiver for the vertical deflection on a scope, and the scanning frequency for the horizontal deection, a pattern is presented which is similar to that lseen in FIG. l0, After an accurate initial adjustment of the zero point, the position of the envelope characteristic represented on the scope can be used as a measurement of the polarization angle a because of the relationship between scanning frequency and envelope. The maximum envelope amplitude indicates a and the minimum envelope amplitude indicates Known techniques may be utilized to determine absolute values of polarization.

With switch 404 in position II, stationary orientation of a is achieved. Any desired D.C. value of a can be adjusted and maintained for an indenite period of time. With switch 404 in position III, a constant rate of change of the polarization vector is achieved. The polarization may be rotated clockwise or counter-clockwise with selectable rotation frequency similar to the operation described for continuous rotation of the antenna pattern in connection with FIG. 8. In the latter case, the sweep signal applied to the horizontal plates `of oscilloscope 508 is either A sin a or A cos a instead of the output Vc from generator 400.

While there has been described what is at present considered to be the preferred embodiment of this invention, it will be obvious to those skilled in the art that various changes and modifications may be made therein without departing from the invention, and it is therefore aimed in the appended claims to cover all such changes and modifications as fall within the true spirit and scope of the invention.

What is claimed is: 1. A steerable antenna system comprising a plurality of antenna elements spaced with respect to a reference antenna element such that a wave front of high frequency radio energy at an antenna element is at a prescribed time phase wit-h respect to the wave front of said high frequency radio energy at said reference antenna, means for generating a signal voltage having a magnitude proportional to the angle of incidence of the wave front of lsaid high frequency radio energy,

discrete phase Shifters in circuit with respective antenna elements except said reference antenna element, said phase Shifters being responsive to said high frequency radio energy,

a prescribed signal source adapted to generate a signal at a given repetition rate,

means responsive to said given repetition rate signal and said signal voltage for generating a control pulse positioned between two successive pulses of said repetition frequency in accordance with the magnitude of said signal voltage,

and discrete means respectively in circuit with said phase shifters and responsive to said control pulse whereby the high frequency radio energy at respective phase Shifters is shifted in phase by an amount corresponding to the time phase of the high frequency radio energy at the associated antenna element.

2. The system in accordance with claim 1 and further including respective adder circuits responsive to the radiofrequency energy outputs of respective phase Shifters and to the radio-frequency energy output of said reference antenna element, said adder circuits being serially connected such that the output of any one phase shifter is added `to the outputs of preceding phase Shifters and the combined added phase shifter outputs are added to the output of the reference antenna element.

3. The system in accordance with claim 1, wherein i3 said antenna elements are linearly spaced along a fixed plane.

4. A steerable antenna system comprising a plurality of antenna elements linearly spaced along a prescribed ground plane with respect to a reference antenna element such that a wave front of high frequency radio energy at an antenna element is at a prescribed time phase angle with respect to the wave front of said high frequency radio energy at said reference antenna, means for generating a signal voltage having a magnitude proportional to the angle of incidence of the wave front of said high frequency radio energy,

`discrete phase Shifters in circuit with respective antenna elements except said reference antenna element,

discrete means respectively in circuit with said phase Shifters and responsive to said signal Voltage for simultaneously generating respective pairs of signals 90 out of phase but functionally related to the time phase angle at the .associated antenna element,

said respective phase Shifters being responsive to the high frequency radio energy derived from `the associated antenna element and its associated signal pair generating means such that the high frequency radio energy at respective phase Shifters is shifted in phase by an vamount corresponding to said time phase angle.

5. A steerable antenna system comprising a plurality of antenna elements llinearly spaced and nonuniformly distributed along a prescribed ground plane with respect to a reference antenna element such that a wave front of high frequency energy at an antenna element is a prescribed time phase angle with respect to the wave front of said high frequency radio energy at said reference antenna,

a prescribed sign-al source adapted to generate a signal pulse at a given repetition rate, means for generating a signal voltage having a magnitude vproportional to the angle of incidence of the wave front of the high frequency radio energy,

means responsive to said given repetition rate signal :and said signal voltage `for generating a control pulse positioned between two successive pulses of said given repetition frequency signal in accordance with the magnitude of said signal voltage,

kdiscrete phase Shifters in circuit with respective antenna elements except said reference antenna element, discrete means respectively in circuit with said phase Shifters and responsive to said control pulse for simultaneously generating respective pairs of signals 90 out of phase but functionally related to the phase angle at the associated antenna element,

said respective phase Shifters being responsive to the high frequency radio energy derived from its associated antenna element and its associated signal pair generating means such that the high frequency Aradio energy at respective phase Shifters is shifted in phase by an amount corresponding to said time phase angle.

6. The system in accordance with claim 5 wherein said respective pairs of signals are sine and cosine functions of the phase angle of the associated antenna element.

7. The system in accordance with claim 5 wherein each of said discrete signal pair generating means comprise a first .and a second auxiliary frequency generator adapted to produce respective wave signals at the same frequency but 90 out of phase, the frequency ratio of the auxiliary frequency and said pulse repetition frequency being a function of the distance in wavelength between the associated antenna element and the reference antenna element.

8. The system in accordance with claim 5 wherein said discrete phase shifters comprise means for generating push-pull high frequency radio energy signals out of phase, a first and second capacitive bridge type modulator,

said Ii-rst modulator being responsive to the zero degree phase high frequency radio signal and the zero phase signal of said signal pair,

said second modulator being responsive to the 90 phase high frequency radio signal and the 90 phase signal of said signal pair.

9. In a steerable antenna system,

a circuit for shifting the phase of a high frequency radio Signal a prescribed time phase angle comprising means for generating two push-pull high frequency radio signals 90 out of phase, a first and second capacitive bridge type modulator respectively connected across the outputs of said push-pull signal generating means,

each of said modulators comprising two voltage variable capacitors and two` fixed capacitors in series connection respectively across the output of said pushpull signal generating means,

means for generating two signals which are functions of said time phase angle and 90 out of phase with respect to each other,

means for combining the zero phase signal function of said time phase angle and the zero phase high frequency signal in said first modulator to produce at the output thereof the product of the combined input signals,

ymeans for combining the 90 phase signal function of said time phase angle and the 90 phase high frequency radio signal in said second modulator to pro duce at the output thereof the product of the combined input signals,

said product signals being combined at a common output connection.

10. The phase Shifter circuit in accordance with claim 9 wherein the two signals which are functions of the time phase angle are applied respectively to the junction of respective serially connected voltage variable capacitors.

11. The system in accordance with claim 4 wherein each phase shifter comprises means for generating two push-pull high frequency radio signals 90 out of phase,

a rst and second capacitive bridge type modulator respectively connected across the outputs of said push-pull signal generating means,

each of said modulators comprising two voltage variable capacitors and two xed capacitors in series connection respectively across the outputs of said push-pull signal generating means,

means for combining the zero phase signal function of said time phase angle and the zero phase high frequency radio signal in said first modulator to produce at the output thereof the product of the combined input signals thereto,

means for combining the 90 phase signal function of said time phase angle and the 90 phase high frequency radio signal in said second modulator to produce at the output thereof the product of the combined input signals thereto,

said product signals being combined at a common output connection.

12. A system for orientating the linear (polarization of an incident wave of high frequency radio energy comprising a pair of quadrature positioned crossed dipole antennas,

one of said antennas being positioned along a prescribed horizontal axis such that the vector representing said incident wave signal forms a prescribed angle of polarization with respect to said horizontal axis, the horizontal and vertical voltage signal components of said incident signal vector being cosine and sine functions, respectively, of said angle of polarization,

a first and second capacitive bridge type modulator responsive, respectively, to said horizontal and vertical voltage signal components,

means for additively combining the outputs of said modulators,

means for generating a signal voltage having a magnitude proportional to said angle of polarization of the incident wave signal,

and means in circuit with said capacitive bridge modulators and responsive to said signal voltage for simultaneously generating a pair of signal voltages which are sine and cosine functions, respectively, of said polarization angle,

said signal voltage cosine function and said horizontal voltage signal component being combined in the rst of said modulators to produce the product of said first modulator combined signals,

said signal voltage sine function and said vertical voltage signal component being combined in the second of said modulators to produce the product of the second modulator combined signals,

the combined output of said modulators being the sum of said product signals. Y

13. A system for orientating the linear polarization of an incident wave of high frequency radio energy cornprising a pair of quadrature positioned crossed dipole antennas,

one of said antennas being positioned along a prescribed horizontal axis such that the vector representing said incident wave signal forms a prescribed angle of polarization with respect to said horizontal axis, the horizontal and vertical voltage signal components of said incident signal vector being cosine and sine functions, respectively, of said angle of polarization,

a iirst and second capacitive bridge type modulator responsive, respectively to said horizontal and vertical voltage signal components,

means for additively combining the outputs of said modulators,

means for generating a signal voltage having a magnitude proportional to the angle of polarization of the incident wave signal,

means for sweeping said signal voltage in accordance with an A.C. scanning signal,

means in circuit with said capacitive bridge modulator and responsive to said swept signal voltage for simultaneously generating a pair of signal voltages which are sine and cosine functions, respectively, of the polarization angle,

said signal voltage cosine function and said horizontal voltage signal component being combined in the first of said modulators to produce the product of said first modulator combined signals,

said signal voltage sine function and said vertical voltage signal component being combined in the second of said modulators to produce the product of said second modulator combined signals,

the combined output of said modulators being the sum of said product signals, a receiver responsive to said combined modulator output signal and adapted to produce an LF signal,

an oscilloscope having vertical and horizontal deflection plates, said vertical deflection plates being responsive to said IF signal, and said horizontal deflection plates being responsive to said scanning signal, the pattern being formed on said oscilloscope face being an indication of said polarization angle.

14. The system in accordance with claim 13 wherein said rst and second modulators each comprise two voltage variable capacitors and two fixed capacitors respectively across the outputs of said quadrature positioned antennas, said signal voltage sine function and signal voltage cosine function being applied to the junction of the variable capacitors in said rst and second modulators respectively.

15. A system for orientating the linear polarization of an incident wave of high frequency radio energy comprising antenna means responsive to the incident signal vector of said high frequency radio energy for producing horizontal and vertical voltage signal components of said incident signal vector which are discrete cosine and sine functions of the angle of polarization of said incident wave measured with reference to a prescribed axis,

a rst and second capacitive bridge type modulator responsive respectively to said horizontal and vertical signal components,

means for additively combining the outputs of said modulators,

means for generating a signal voltage having a magnitude proportional to said angle of polarization of the incident Wave signal,

and means in circuit with said capacitive bridge modulators and responsive to said signal voltage for simul. taneously generating a pair of signal voltages which are sine and cosine functions, respectively, of said polarization angle,

said signal voltage cosine function and said horizontal voltage signal component being combined in the irst of said modulators to produce the product of said iirst modulator combined signals,

said signal voltage sine function and said vertical volttage function signal component being combined in the second of said modulators to produce the product of the second modulator combined signals,

the combined output of said modulators being the sum of said product signals.

References Cited bythe Examiner UNITED STATES PATENTS 2,245,660 6/1941 Feldman et al 343--100 X 2,464,276 3/ 1949 Varian. 3,036,210 5/1962 Lehan et al. 343-100 CHESTER L. JUSTUS, Primary Examiner.

T. H. TUBBESING, H. C. WAMSLEY,

Assistant Examiners. 

4. A STEERABLE ANTENNA SYSTEM COMPRISING A PLURALITY OF ANTENNA ELEMENTS LINEARLY SPACED ALONG A PRESCRIBED GROUND PLANE WITH RESPECT TO A REFERENCE ANTENNA ELEMENT SUCH THAT A WAVE FRONT OF HIGH FREQUENCY RADIO ENERGY AT AN ANTENNA ELEMENT IS AT A PRESCRIBED TIME PHASE ANGLE WITH RESPECT TO THE WAVE FRONT OF SAID HIGH FREQUENCY RADIO ENERGY AT SAID REFERENCE ANTANNA, MEANS FOR GENERATING A SIGNAL VOLTAGE HAVING A MAGNITUDE PROPORTIONAL TO THE ANGLE OF INCIDENCE OF THE WAVE FRONT OF SAID HIGH FREQUENCY RADIO ENERGY, DISCRETE PHASE SHIFTERS IN CIRCUIT WITH RESPECTIVE ANTENNA ELEMENTS EXCEPT SAID REFERENCE ANTENNA ELEMENT, DISCRETE MEANS RESPECTIVELY TO SAID SIGNAL VOLTAGE FOR SHIFTERS AND RESPONSIVE TO SAID SIGNAL VOLTAGE FOR SIMULTANEOUSLY GENERATING RESPECTIVE PAIRS OF SIGNALS 90* OUT OF PHASE BUT FUNCTIONALLY RELATED TO THE TIME PHASE ANGLE AT THE ASSOCIATED ANTENNA ELEMENT, SAID RESPECTIVE PHASE SHIFTERS BEING RESPONSIVE TO THE HIGH FREQUENCY RADIO ENERGY DERIVED FROM THE ASSOCIATED ANTENNA ELEMENT AND ITS ASSOCIATED SIGNAL PAIR GENERATING MEANS SUCH THAT THE HIGH FREQUENCY RADIO ENERGY AT RESPECTIVE PHASE SHIFTERS IS SHIFTED IN PHASE BY AN AMOUNT CORRESPONDING TO SAID TIME PHASE ANGLE. 